Ic op amp cookbook pdf free download

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Books for People with Print Disabilities. Internet Archive Books. American Libraries. Moreover, aspects of the practical realization have been significantly expanded with complete design examples and with typical building blocks for control systems. The book is ideal for upper level students in electrical and mechanical engineering, for whom a course in Feedback Controls is usually required.

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Electroanalytical Chemistry Author : Gary A. It provides insight beyond the field of analysis as students address problems arising in many areas of science and technology.

The book also emphasizes electrochemical phenomena and conceptual models to help readers understand the influence of experimental conditions and the interpretation of results for common potentiometric and voltammetric methods. Electroanalytical Chemistry: Principles, Best Practices, and Case Studies begins by introducing some basic concepts in electrical phenomena. It then moves on to a chapter that examines the potentiometry of oxidation-reduction processes, followed by another on the potentiometry of ion selective electrodes.

Other sections look at: applications of ion selective electrodes; controlled potential methods; case studies in controlled potential methods; and instrumentation. Introduces the principles of modern electrochemical sensors and instrumental chemical analysis using potentiometric and voltammetric methods Develops conceptual models underlying electrochemical phenomena and useful equations Illustrates best practice with short case studies of organic reaction mechanisms using voltammetry and quantitative analysis with ion selective electrodes Offers instructors the opportunity to select focus areas and tailor the book to their cours.

This book begins with an overview of embedded systems and microcontrollers followed by a comprehensive in-depth look at the MSP The coverage included a tour of the microcontroller's architecture and functionality along with a review of the development environment. Start using the MSP armed with a complete understanding of the microcontroller and what you need to get the microcontroller up and running!

Details C and assembly language for the MSP Companion Web site contains a development kit Full coverage is given to the MSP instruction set, and sigma-delta analog-digital converters and timers. The Hands-on XBee Lab Manual takes the reader through a range of experiments, using a hands-on approach. Each section demonstrates module set up and configuration, explores module functions and capabilities, and, where applicable, introduces the necessary microcontrollers and software to control and communicate with the modules.

Experiments cover simple setup of modules, establishing a network of modules, identifying modules in the network, and some sensor-interface designs. This book explains, in practical terms, the basic capabilities and potential uses of XBee modules, and gives engineers the know-how that they need to apply the technology to their networks and embedded systems. Titus is the inventor of the first personal-computer kit, the Mark-8, now in the collection at the Smithsonian Institution.

Both RTI and RTO noise errors are calculated the same way as offset errors, except that the noise of two sections adds as the root mean square. Of particular concern are situations in which signal transmission lines are long and signal strength is low. This is the classic application for an in-amp, since its inherent common-mode rejection allows the device to extract weak differential signals riding on strong common-mode noise and interference.

One potential problem that is frequently overlooked, however, is that of radio frequency rectification inside the in-amp. When strong RF interference is present, it may become rectified by the IC and then appear as a dc output offset error.

Unfortunately, RF rectification occurs because even the best in-amps have virtually no common-mode rejection at frequencies above 20 kHz.

Once rectified, no amount of low-pass filtering at the in-amp output will remove the error. If the RF interference is of an intermittent nature, this can lead to measurement errors that go undetected. The filter needs to do three things: remove as much RF energy from the input lines as possible, preserve the ac signal balance between each line and ground common , and maintain a high enough input impedance over the measurement bandwidth to avoid loading the signal source.

Figure is a simplified version of the RFI circuit. Therefore, resistors R1a and R1b and capacitors C1a and C1b should always be equal. As shown, C2 is connected across the bridge output so that C2 is effectively in parallel with the series combination of C1a and C1b.

Thus connected, C2 very effectively reduces any ac CMR errors due to mismatching. Note that the filter does not affect dc CMR. The RFI filter has two different bandwidths: differential and common mode. This RC time constant is established by the sum of the two equal-value input resistors R1a, R1b , together with the differential capacitance, which is C2 in parallel with the series combination of C1a and C1b.

At unity gain, there was no measurable dc offset shift. All component leads should be made as short as possible. The input filter common should be connected to the amplifier common using the most direct path.

Avoid building the filter and the in-amp circuits on separate boards or in separate enclosures, as this extra lead length can create a loop antenna. A further precaution is to use good quality resistors that are both noninductive and nonthermal low TC. However, all three capacitors need to be reasonably high Q, low loss components. First, decide on the value of the two series resistors while ensuring that the previous circuitry can adequately drive this impedance.

A differential bandwidth of 10 times the highest signal frequency is usually adequate. Then select values for capacitors C1a and C1b, which set the common-mode bandwidth. Accordingly, the same input resistors were used, but capacitor C2 was increased approximately five times to 0. Note that this increased bandwidth does not come free.

It requires the circuitry preceding the in-amp to drive a lower impedance load and results in somewhat less input overload protection. A micropower in-amp, such as the AD, with its low input stage operating current, is a good example. The filter bandwidth is approximately Hz. RFI circuit for AD series in-amp. RFI suppression circuit for the AD With the values shown, the bandwidth of this filter is approximately Hz. Figure shows the recommended RFI filter for this in-amp.

Using the filter, there was no measurable dc offset error. AD RFI suppression circuit. AD RFI filter circuit. These built-in filters on the pins limit the interference bandwidth and provide good RFI suppression without reducing performance within the pass band of the in-amp. Using the component values shown, this filter has a common-mode bandwidth of approximately 40 kHz. Signals present at this pin appear as common-mode signals and create problems. Furthermore, additional filtering at the VPOS and VNEG pins should provide better reduction of unwanted behavior compared with filtering at the other pins.

These are very small, three terminal devices with four external connections—A, B, G1, and G2. The G1 and G2 terminals connect internally within the device. The internal plate structure of the X2Y capacitor forms an integrated circuit with very interesting properties.

Electrostatically, the three electrical nodes form two capacitors that share the G1 and G2 terminals. The manufacturing process automatically matches both capacitors very closely. This usually allows the omission of capacitor C2, with subsequent savings in cost and board space. X2Y electrostatic model. RF attenuation, X2Y vs. Figure is a graph contrasting the RF attenuation provided by these two filters.

Common-mode filter using X2Y capacitor. Conventional RC common-mode filter. For a full listing of X2Y manufacturers, visit www. A common-mode choke is a two-winding RF choke using a common core. Any RF signals that are common to both inputs will be attenuated by the choke. The common-mode choke provides a simple means for reducing RFI with a minimum of components and provides a greater signal pass band, but the effectiveness of this method depends on the quality of the particular common-mode choke being used.

A choke with good internal matching is preferred. Another potential problem with using the choke is that there is no increase in input protection as is provided by the RC RFI filters. Using an AD in-amp with the RF choke specified, at a gain of , and a 1 V p-p common-mode sine wave applied to the input, the circuit of Figure reduces the dc offset shift to less than 4.

The high frequency common-mode rejection ratio was also greatly improved, as shown in Table AC CMR vs. In these cases, an RC input filter or an X2Y-based filter is a better choice. To test these circuits for RFI suppression, connect the two input terminals together using very short leads.

Using an oscilloscope, adjust the generator for a 1 V peak-to-peak output at the generator end of the cable. Set the in-amp to operate at high gain such as a gain of For measuring high frequency CMR, use an oscilloscope connected to the in-amp output by a compensated scope probe and measure the peak-to-peak output voltage i.

When calculating CMRR vs. In some cases, band-pass filtering reducing response both below and above the signal frequency can be employed for an even greater improvement in measurement resolution. This filter provides high dc precision at low cost while requiring a minimum number of components. Furthermore, since the input bias current of these op amps is as low as their input offset currents over most of the MIL temperature range, there is rarely a need to use the normal balancing resistor along with its noisereducing bypass capacitor.

Note that component values can simply be scaled to provide corner frequencies other than 1 Hz see Table If a 2-pole filter is preferred, simply take the output from the first op amp. Specified values are for a —3 dB point of 1. For other frequencies, simply scale capacitors C1 through C4 directly; i. Thus, lower capacitor values may be used, reducing cost and space. A 4-pole low-pass filter for data acquisition. The input frequency used should be somewhat lower than the —3 dB bandwidth of the circuit.

When a very high speed, wide bandwidth in-amp is needed, one common approach is to use several op amps or a combination of op amps and a high bandwidth subtractor amplifier. These discrete designs may be readily tuned up for best CMR performance by external trimming.

A typical circuit is shown in Figure The input amplitude should be set at 20 V p-p with the inputs tied together. The ac CMR trimmer is then nulled-set to provide the lowest output possible. If the best possible settling time is needed, the ac CMR trimmer may be used, while observing the output wave form on an oscilloscope.

Note that, in some cases, there will be a compromise between the best CMR and the fastest settling time. The voltage should be adjusted to provide a 10 V dc input.

A dc CMR trimming potentiometer would then be adjusted so that the outputs are equal and as low as possible, with both a positive and a negative dc voltage applied.

External dc and ac CMR trim circuit for a discrete 3-op amp in-amp. This would produce a common-mode output voltage of half the ADC reference voltage.

Note that VREF2 is high impedance but cannot swing to the supply rails of the part. One very common application sets the common-mode output voltage to the midscale of a differential ADC. Differential output in-amp circuit. Figure shows two conventional methods used to measure a large signal. One comprises a 2-resistor divider and an output buffer, the other an inverter with a large value input resistor. Both approaches suffer from the fact that only one resistor dissipates power and, therefore, is self-heating and the change in resistance due to temperature change results in a large nonlinearity error.

Another problem associated with these approaches concerns the amplifier: The combination of offset current, offset voltage, CMRR, gain error, and drifts of the amplifier and resistors may significantly reduce the overall system performance. Two conventional methods of measuring high voltage. Figure is a schematic of a circuit that can measure in excess of volts peak-to-peak input with less than five parts per million nonlinearity error.

The circuit attenuates an input signal 20 times and also provides output buffering. The amplifier, as well as the attenuator resistors, are all packaged together inside the AD IC so that both resistors in the attenuator string are at the same temperature.

The amplifier stage employs superbeta transistors, so that offset current error and bias current errors are small. Also, since there is no noise gain i. Performance photo: top, input voltage V p-p , bottom, output voltage 20 V p-p. New high voltage measurement system. A 30 pF capacitor adds a pole and a zero to the feedback gain, so stability is maintained and the system bandwidth is maximized. Cross plot of the high voltage measurement system. Figure is a performance photo showing the output at 5 V p-p per division vs.

Figure , also a cross plot, shows nonlinearity vs. Consequently, the voltage drop across R2 is set by VIN, according to the following equation: Figure shows the AD configured as an integrator. This configuration can be used within a PID proportional integral differential loop in a control system.

The input offset voltage of this configuration will be proportional to the size of R1, assuming that the system is in a steady state condition. Therefore, the input offset voltage of the integrator is determined primarily by the size of R1 along with the internal offsets of the AD, assuming that the system is in a steady state condition. Precision 61 mA dc current source. The value of this current can range between 61 mA.

This is 0. Low frequency differential input integrators for PID loop. It features an extended frequency range over which the instrumentation amplifier has good common-mode rejection Figure The circuit consists of three instrumentation amplifiers. Two of these, U1 and U2, are correlated to one another and connected in antiphase. It is not necessary to match these devices because they are correlated by design. The overall gain of the system can be determined by adding external resistors.

Without any external resistors, the system gain is 2 Figure The performance of the circuit with a gain of is shown in Figure A composite instrumentation amplifier. CMR of the system at a gain of Since U1 and U2 are correlated, their common-mode errors are the same. Therefore, these errors appear as common-mode input signal to U3, which rejects them. The differential-input stage of the ADC normally will reject the common-mode signal.

CMR of the circuit in Figure at 20 kHz. The output signal is devoid of all dc errors associated with the in-amp and the detector, including offset and offset drift. Similarly, the signal at the input of the AD is ac; the signal is dc at the end of the low-pass filter following the AD Ultimately, a precision dc signal is obtained.

In contrast, the AD continues to reject common-mode signals beyond 10 kHz. One solution is to use an ac signal to excite the bridge, as shown in Figure What results If an ac source is not available, a commutating voltage may be constructed using switches. Using an ac signal to excite the bridge. Specifically designed for use ahead of an analog-to-digital converter, the AD is extremely useful as an input scaling and buffering amplifier. The AD uses an absolute minimum of external components.

Its tiny MSOP provides these functions in the smallest size package available on the market. Real-world measurement requires extracting weak signals from noisy sources. Even when a differential measurement is made, high common-mode voltages are often present. The usual solution is to use an op amp or, better still, an in-amp, and then perform some type of low-pass filtering to reduce the background noise level. When used with a differential signal source, an in-amp circuit using a monolithic IC will improve common-mode rejection.

However, signal sources greater than the power supply voltage, or signals riding on high common-mode voltages, cannot handle standard in-amps. In addition, in-amps using a single external gain resistor suffer from gain drift. Finally, lowpass filtering usually requires the addition of a separate op amp, along with several external components.

This drains valuable board space. The AD eliminates these common problems by functioning as a scaling amplifier between the sensor, the shunt resistor, or another point of data acquisition, as well as the ADC. Its V max input range permits the direct measurement of large signals or small signals riding on large common-mode voltages.

Pin 3 is grounded, thus operating amplifier A1 at a gain of 0. Basic differential input connection with single-pole LP filter. Pin 4 allows connecting an external capacitor to this point, providing single-pole low-pass filtering. Changing the Output Scale Factor Figure reveals that the output scale factor of the AD may be set by changing the gain of amplifier A2.

This uncommitted op amp may be operated at any convenient gain higher than unity. When configured, the AD may be set to provide circuit gains between 0. Since the gain of A1 is 0. This connection is the same as the basic wide input range circuit of Figure , but with Pins 5 and 6 strapped, and with an external resistor, RG, connection between Pin 4 and ground.

The pin strapping operates amplifier A2 at unity gain. The gain for this connection equals 0. AD connection for gains less than 0. Differential input circuit with 2-pole low-pass filtering. The second pole is created by an external RC time constant in the feedback path of A2, consisting of capacitor C2 across resistor R F.

Here, 2-pole low-pass filtering is added for the price of one additional capacitor C2. Frequency response of the 2-pole low-pass filter.

The AD provides this complete function using the smallest IC package available. Since all resistors are internal to the AD gain block, both accuracy and drift are excellent. The gain will also vary with temperature because each resistor will drift differently. Monolithic resistor networks can be used for precise gain setting, but these components increase cost, complexity, and board space.

Values have been rounded off to match standard resistor and capacitor values. Capacitors C1 and C2 need to be high Q, low drift devices; low grade disc ceramics should be avoided. High quality NPO ceramic, Mylar, or polyester film capacitors are recommended for the lowest drift and best settling time. Unfortunately, All of these pin-strapped circuits using no external components have a gain accuracy better than 0.

The gain block may be configured to provide different gains by strapping or grounding the appropriate pin. The gain block itself consists of two internal amplifiers: a gain of 0. Therefore, the positive input of A2 will be forced by feedback from the output of A2 to be 0 V as well.

The output of A1 then must also be at 0 V. With Pin 3 grounded, the positive input of A1 is at 0 V, so feedback will force the negative input of A1 to zero as well.

The two connections will have different input impedances. The —3 dB bandwidth for both circuits is approximately kHz for 10 mV and 95 kHz for mV input signals. Companion circuit providing a gain of — Note that this series resistance is negligible compared to the very high input impedance of amplifier A1. The gain from Pin 8 to the output of A1 is 0.

Therefore, feedback will force the output of A2 to equal 10 VIN. The —3 dB bandwidth of this circuit is approximately kHz for 10 mV and 95 kHz for mV input signals. The input signal is applied between the V REF pin and ground. Therefore, the voltage at the output of A1 will equal V IN 1. Increased BW Gain Block of —9. The output of amplifier A1 feeds back its positive input by connecting Pin 4 and Pin 1 together.

Precision —10 gain block with feedforward. The advantage of such a system is its ability to operate with two remotely connected power supplies, even if their grounds are not the same.

In such cases, it is necessary for the output to be linear with respect to the input signal, and any interference between the grounds must be rejected. Figure shows such a circuit. The common-mode signal, VCM, will be rejected. In order to reduce the voltage at Pin 6, an inverter with a gain of 9 is connected between Pin 6 and its reference. The inverter sets the gain of the transmitter such that for a 10 V input, the voltage at Pin 6 only changes by 1 V; yet, the difference between Pin 6 and its reference is 10 V.

Since the gain between the noninverting terminal of the OP27 and the output of the AD is 1, no modulation of the output current will take place as a function of the output voltage VOUT. Current transmitter. Figure is the transfer function of the output voltage VOUT vs. Figure is a demonstration of how well the transmitter rejects ground noise. Interference rejection. Transfer function. As shown, two AD difference amplifiers are connected in antiphase.

The difference signal is amplified by 1. The benefits of this configuration go beyond simply interfacing with the ADC. The waveforms show a 10 V input signal top , the signals at the output of each AD middle , and the differential output bottom.

The common-mode input top measures 20 V p-p. The AD easily rejects residual common-mode signals at the output of the difference amplifiers. Figure shows the commonmode error at the output of the AD It is a variation of an inverting amplifier. Point X is a virtual ground and referred to as a summing junction. A traditional summing amplifier. Furthermore, this circuit has many disadvantages, such as a low input impedance, plus different impedances for positive and negative inputs.

It requires low bandwidth, and highly matched resistors are needed. Figure is the schematic of a high speed summing amplifier, which can sum up as many as four input voltages without the need for an inverter to change the sign of the output. This could prove very useful in audio and video applications. The circuit contains three, low cost, high speed instrumentation amplifiers. The inputs are very high impedance, and the signal that appears at the network output is noninverting.

VO R2 V2 This indicates that the output is a weighted sum of the inputs, with the weights being determined by the resistance ratio. If all resistances are equal, the circuit yields the inverted sum of its inputs. A summing circuit with high input impedance. The top trace is the input signal for all four inputs.

The middle trace is the sum of inputs V1 and V2. The bottom trace is the output of the system, which is the total sum of all four inputs. As we can see, the —3 dB point is about MHz. High voltage monitor. VOUT, the integrator output and the measurement output, sources the required current to maintain the common-mode voltage.

R1 and C1 compensate the system to a bandwidth of kHz. For example, a V p-p input signal will produce a 21 V p-p output. Performance photo of the circuit in Figure Frequency response of summing circuit in Figure High common-mode voltage difference amplifiers have been used to monitor current. However, these versatile components can also be used as voltage dividers, enabling remote monitoring of voltage levels as well.

Figure shows a precision monitor using just two integrated circuits that derives its power from the —48 V supply. A low cost transistor and Zener diode combination provide 15 V supply voltage for the amplifiers.

Connected as shown, it reduces the differential input voltage by approximately 19 V, thus acting as a precision voltage divider. An additional amplifier is required for loop stability. Nonlinearity vs. The batteries provide backup power during ac power line outages and help regulate the 48 V dc supply voltage. Although nominally —48 V, the dc voltage on the telephone lines can vary anywhere from —40 V to —80 V and is subject to surges and fluctuations. Linearity errors from —40 V to —80 V are nearly immeasurable.

Figures and are linearity and temperature drift curves for this circuit. Precision remote voltage measurement of —48 V power distribution bus. Typical applications include hydraulic transmission control and diesel injection control. Output vs. When the switch is opened, the voltage reversal across the inductive load causes the commonmode voltage to be held one diode drop above the battery by the clamp diode.

An inductive load solenoid is tied to a power supply. An advantage of placing the shunt on the high side is that the entire current, including the recirculation current, can be measured, since the shunt remains in the loop when the switch is off. In addition, diagnostics can be enhanced because shorts to ground can be detected with the shunt on the high side. Temperature drift of the 48 V bus monitor.

Low-side switch. In this case, both the switch and the shunt are on the high side. When the switch is off, this removes the battery from the load, which prevents damage from potential shorts to ground while still allowing the recirculating current to be measured and providing for diagnostics. Removing the power supply from the load for the majority of the time minimizes the corrosive effects that could be caused by the differential voltage between the load and ground.

When using a high-side switch, the battery voltage is connected to the load when the switch is closed, causing the common-mode voltage to increase to the battery voltage. In this case, when the switch is opened, the voltage reversal across the inductive load causes the common-mode voltage to be held one diode drop below ground by the clamp diode.

Motor control application. The AD measures current in both directions as the H-bridge switches and the motor changes direction. The output of the AD is configured in an external reference bidirectional mode.

High-side switch. Motor Control A typical application for the AD is as part of the control loop in H-bridge motor control. In this case, the AD is placed in the middle of the H-bridge see Figure so that it can accurately measure current in both directions by using the shunt available at the motor. Figure shows the AD configured to amplify the signal from a classic resistive bridge. This circuit will work in either dual- or single-supply mode.

Typically, the bridge will be excited by the same voltage used to power the in-amp. Connecting the bottom of the bridge to the negative supply of the in-amp usually either 0, —5 V, —12 V, or —15 V sets up an input common-mode voltage that is optimally located midway between the supply voltages.

It is also appropriate to set the voltage on the REF pin to midway between the supplies, especially if the input signal will be bipolar. This instability in the ground reference causes the measurements that could be made with a simple ground referenced op amp to be inaccurate.

A classic bridge circuit for low power applications. This corresponds to the input range of the ADC. Figure shows one-half of the AD being used to buffer the AD, a 1. The other half of the AD is configured as a unity-gain inverter and generates the other bridge input of —4. Resistors R1 and R2 provide a constant current for bridge excitation. The AD low power instrumentation amplifier is used to condition the differential output voltage of the bridge.

A single-supply data acquisition system. Low dropout bipolar bridge driver. High quality transducers typically provide a highly linear output, but at a very low level and a characteristically high output impedance. Furthermore, the high output impedance of the typical transducer may require that the in-amp have a low input bias current. Table gives typical characteristics for some common transducer types.

Since most transducers are slow, bandwidth requirements of the in-amp are modest: A 1 MHz small signal bandwidth at unity gain is adequate for most applications. Potentials measured on the skin range from 0. The AD solves many of the typical challenges of measuring these body surface potentials. Its rail-to-rail output offers wide dynamic range allowing for higher gains than would be possible using other instrumentation amplifiers.

JFET inputs offer a large input capacitance of 5 pF. In addition, the AD JFET inputs have ultralow input bias current and no current noise, making it useful for ECG applications where there are often large impedances.

Figure shows an example of an ECG schematic. Following the AD is a 0. An active, fifth-order, low-pass Bessel filter removes signals greater than approximately Hz. An OP buffers, inverts, and gains the common-mode voltage taken at the midpoint of the AD gain setting resistors. This right leg drive circuit helps cancel common-mode signals by inverting the common-mode signal and driving it back into the body. A kV series resistor at the output of the OP limits the current driven into the body.

An example of an ECG schematic. Each in-amp is followed by a high-pass filter that removes the dc component from the signal. It is common practice to omit one of the in-amps and determine the third output by software or hardware calculation. So, unshielded twisted pair cable is recommended for this circuit. For low speed applications that require driving long lengths of shielded cable, the AMP01 should be substituted for the AMP03 device. Proper safeguards, such as isolation, must be added to this circuit to protect the patient from possible harm.

The AD operates with full accuracy on standard 5 V power supply voltages. If heavy output currents are expected, and there is a need to sense a load that is some distance away from the circuit, voltage drops due to trace or wire resistance can cause errors.

It makes use of the large common-mode range of the AD The current being measured is sensed across resistor RS. The sense terminal completes the feedback path for the instrumentation amplifier output stage and is normally connected directly to the in-amp output. This connection puts the IR drops inside the feedback loop of the in-amp and virtually eliminates any IR errors.

This circuit will provide a 3 dB bandwidth better than 3 MHz. A remote load sensing connection. Current sensor interface. The addition of the AD isolates the in-amp from the load, thus greatly reducing any thermal effects.

Output buffer for low power in-amps. The signal from a 4 to 20 mA transducer is single-ended. This initially suggests the need for a simple shunt resistor to convert the current to a voltage at the high impedance analog input of the converter. However, any line resistance in the return path to the transducer will add a currentdependent offset error.

So, the current must be sensed differentially. In this example, a With no gain resistor present, the AD amplifies the mV input voltage by a factor of 5 to 2. The zero current of 4 mA corresponds to a code of , and the LSB size is 0. Figure shows a thermocouple application where one side of the J-type thermocouple is grounded.

Table lists some of these products. A 4 to 20 mA receiver circuit. A thermocouple amplifier using a low power, single-supply in-amp. In an ADC, the available resolution equals 2n — 1, where n is the number of bits. For example, an 8-bit converter provides a resolution of 28 — 1, which equals In this case, the full-scale input range of the converter divided by will equal the smallest signal it can resolve. For example, an 8-bit ADC with a 5 V full-scale input range will have a limiting resolution of Table provides input resolution and full-scale input range using an ADC with or without an in-amp preamplifier.

Note that the system resolution specified in the figure refers to that provided by the converter together with the in-amp preamp if used. Also, note that for any low level measurement, not only are low noise semiconductor devices needed, but also careful attention to component layout, grounding, power supply bypassing, and often, the use of balanced, shielded inputs. For many applications, an 8-bit or bit converter is appropriate. The decision to use a high resolution converter alone, or to use a gain stage ahead of a lower resolution converter, depends on which is more important: component cost, or parts count and ease of assembly.

One very effective way to raise system resolution is to amplify the signal first, to allow full use of the dynamic range of the ADC. However, this added gain ahead of the converter will also increase noise. Therefore, it is often useful to add low-pass filtering between the output of an in-amp or other gain stage and the input of the converter.

Also, in most cases, the system bandwidth should not be set higher than that required to accurately measure the signal of interest.

A good rule of thumb is to set the —3 dB corner frequency of the low-pass filter at 10 to 20 times the highest frequency that will be measured. For example, using an in-amp with a gain of 10 ahead of an 8-bit, 5 V ADC will increase circuit resolution from Typical System Resolutions vs.